1、反激變換器的例子Analysisof basicwaveforms基本波形分析The analysisof the basic waveformswill be done on a simula ted exampleof a flyback converteroperating in discontinuouscond uction mode.Typical drain-sourcevoltage waveform of the primary side switch is shown in Fig. 16.在電感電流斷續模式下運行的反激變換器的典型一次側 漏源極開關電壓波形見圖1 6 o&

2、gt;善s與30Fig. 16 Typical drain-sourcevoltage of the MOSFET in a flyback圖1 6反激變換器的典型漏源極電壓Thesedrain-sourcevoltage waveformscan be theoretically distin guishedinto typical elementsDifferent physical phenomenanfluen ce the waveform at given time interval. Fig. 17 and Tab. 4 demo nstrate the main element

3、sof the voltage waveform. The superposit ion of all theseelementsresults in a typical drain-sourcevoltage s hown in Fig. 16.這些漏源極電壓波形能用典型的理論來描述。各個時間段有不同物理現象影響這些波形。圖1 7和平臺4描述了電壓 波形的主要原理。把這些原理按時序整合呈現由圖1 6所示 的典型漏源極電壓。Fig. 17 Main elementsof the drain-sourcevoltage圖1 7漏源極電壓的主要原理Element 1:voltage fall du

4、ring turning on原理1 :開通期間的電壓下降過程Element 2:parasitic oscillation during turning on due to current spike2011-io-20105 10原理2:在開通期間因寄生震蕩產生的電流尖刺Element 3:voltage rise during turning off原理3 :關斷期間的電壓上升Element 4: clamping voltage of snubberso06040205.6 10 '5.S -10原理4:緩沖電路的鉗位電壓Element 5:parasitic oscillat

5、ion after clamping involving mainly the output capacitance of the MOSFET and the leakage inductance of the transformer原理5:鉗位過程結束后主要由場效應晶體管輸由電容和變壓器漏感引起的寄生振蕩500-502.S3 io-6 3.2 io-*1-100Element 6:parasitic oscillation after flyback phase involving mainly the output capacitance of the MOSFET and the ma

6、gnetization inductance of the transformer100原理6:磁芯存儲磁能釋放完畢后主要由場效應晶體管輸由電容和變壓器電感引起的寄生振蕩EiemenlZ：reflected voltage during Hie fl/back phaseElement a：main rectangular signal with bus ampfitude, $府"夕 msuf ： i*原理7:反激變換器釋放磁能期間的反射電壓原理8:與直流母線電壓等幅的主要方波Tab. 4 Main elementsof the drain-sourcevoltage平臺4漏源極電

7、壓的主要原理The spectrumof the whole drain-sourcewaveform (Fig. 16) is presentedn Fig. 18.圖1 6所示的漏源極電壓呈現的電磁干擾頻譜見圖1 8 ofrequency. Hz9SFFig. 18 Spectrumof the drain-sourcevoltage (as shown in Fig. 16)圖1 8 圖1 6所示的漏源極電壓呈現的電磁干擾頻譜The spectraof the main elementsof the drain-sourcevoltage c an be found in Fig. 20

8、. Fig. 19 is exactly the sameas Fig. 17 and has been repeatedhere for better under-standing.圖2。描述了漏源極電壓主要原理產生的電磁干擾頻譜。 為便于理解，將圖1 7映射成圖1 9。Fig. 19 Main elementsof the drain-sourcevoltage (repeatedsameas Fig. 17)圖1 9漏源極電壓的主要原理(正確重復 圖1 7 )Fig. 20 Spectraof the main elementsof the drain-sourcevoltage圖2 0

9、漏源極電壓主要原理產生的電磁干擾頻譜This method allows associatingcertain parts of the spectrum with their root causesj.e. the peak at 20 MHz in the spectrum of the drain-sourcevoltage is causedby the parasitic oscillation d ue to the output capacitanceof the MOSFET and the leakagein ductanceof the transformer.這種方法可以

10、確定電磁干擾頻譜中某些頻點的來源，也就是說漏源極電壓產生的電磁干擾頻譜中的2。兆赫茲峰點 是鉗位過程結束后主要由場效應晶體管輸由電容和變壓器 漏感引起的寄生振蕩產生的。The analysisof the drain current of the primary switch will b e donein the sameway. Fig. 21 demonstrates typical drain cur rent in a DCM flyback.對一次側開關的漏極電流進行分析采用相同的方法。圖21展示由一個工作于電感電流斷續模式反激變換器的典型漏極電流。Fig. 21 Typical

11、drain current in a flyback圖2 1反激變換器的典型漏極電流This waveform can be presentedas a superpositionof the following elements(Fig. 22 and Tab. 5). The superpositionof all theseelementsresults in a typical drain current shown in Fig. 21.這個波形可以被看作是下列原理的疊加(圖2 2和平臺5) o全部這些波形的疊加整合結果變成圖2 1所示的典型 漏極電流。amWi i ssT®

12、; trrrw £. JiHflP13PFig. 22 Main elementsof the drain current圖2 2漏極電流的主要原理Element V main trianale of the drain current原理1 ：漏極電流的主要三角波形Element 2:current spike during turning on due to parasitic capacitances of the circuit原理2:在開關開通期間因寄生分布電容引起的電流尖刺Element 3:parasitic oscillation after clamping inv

13、olving mainly the output capacitance of the MOSFET and the leakage inductance of the transformer原理3:鉗位過程結束后主要由場效應晶體管輸由電容和變壓器漏感引起的寄生振蕩Element 4: parasitic oscillation after flyback phase involving mainly the output capacitance of the MOSFET and the原理4:磁芯存儲磁能釋放完畢后主要由場效應晶體管輸由電容和變壓器電感引起的寄生振蕩Tab. 5 Main

14、elementsof the drain current平臺5漏極電流的主要原理The spectrumof the whole drain current waveform (Fig. 21) is presentedn Fig. 23.全部漏極電流波形產生的電磁干擾頻譜（圖2 1 ）呈現在圖2 3。Fig. 23 Spectrumof the drain current (as shown in Fig. 22)flIDw 腫 口 4圖2 3漏極電流產生的電磁干擾頻譜(與圖2 2相同)The spectraof the main elementsof the drain current c

15、an be found in Fig. 25. Fig. 24 is exactly the sameas Fig. 22 and ha s beenrepeatedfor better understanding.漏極電流主要原理產生的電磁干擾頻譜見圖2 5。圖2 4和圖2 2相同i hT*i.s -w-11 tiimp £ ?-ifl_*h i.Fig. 24 Main elementsof the drain current圖2 4漏極電流的主要原理Fig. 25 Spectraof the main elementsof the drain current 圖2 5漏極電流主

16、要原理產生的電磁干擾頻譜As in caseof drain-sourcevoltage this methodallows to associ ate the elementsof the drain current waveform with its contributi on to the whole spectrum.For example,the peak at 20 MHz inthe spectrumis causedby the parasitic oscillation due to the out put capacitanceof the MOSFET and the l

17、eakageinductanceof t he transformer.就象漏源極電壓的例子那樣，用這種方法也可以找由漏極電流的哪一部分對電磁干擾頻譜產生影響。舉例說明，2 0 兆赫茲的峰點是鉗位過程結束后主要由場效應晶體管輸由 電容和變壓器漏感引起的寄生振蕩產生的。This methodof separatingthe waveform in time domain into i ts main elementshelps to find out what part of the spectrumin frequencydomain causedby what related physica

18、lphenomenaTh e separationinto main elementsshould be done in respectof reaso nable eventsin the power circuit like on and off slopes,oscillatio ns, clamping,snubbering,reflectedvoltage, etc.這種在時域里對主要原理進行拆分的方法有助于我生產 生電磁干擾頻段的干擾源。這種離析主要原理的手法有助于 合理審視電源電路里諸如變化速率、振蕩、鉗位、緩沖、反 射電壓等過程。In this flyback exampleo

19、nly the primary switch has beenanal yzed as active sourceof electrical noise.There are also others,like secondaryside diodes or synchronousectifier, control IC (especiall y its gate drive), etc. In order to obtain morecompleteanalysisal l theseinterferencesourceshave to be analyzed.在這個反激變換器里只對一次側開關進

20、行電磁噪聲產生 的分析。但是還有其他的部分，象二次側的二極管或同步整 流器、控制集成電路（尤其是它們的柵極驅動）等等。按順 序分析將獲得更完善的關于這些電磁干擾源的解析。However,it is impossibleto predict the conductedEMI spectr um using this approachdue to the fact, that only interferencesou rces are consideredThereis no analysis of the spreadingpaths of the interferencein this met

21、hod.然而，這種方法不可能預知用頻譜反映的電磁干擾的實際 行為，僅僅是干擾源被重視起來。在那里沒有對分布參數產 生的干擾進行分析的方法。Neverthelessthe associationof harmonicsroot causewith the respectecphysical phenomenavill reducethe efforts of EMI reduc tion. The impact of the identified root causecan be reducednot only by filtering, but also by meansof influenc

22、ing the root cause itself.不過，重視物理現象并不能成就電磁干擾的降低。降低干擾并不僅僅是濾波，也同樣意味著干擾源自身的影響。Operationmodesof discontinuouslyback converter電感電流斷續工作反激式變換器的運行模式The flyback converterrunning in discontinuousconductionmode can be operatedin hard switching or quasi resonant (or valley switching, or ZVS) moderegardingthe pr

23、imary side switch. The differencebetween a hard switching and quasi resonantflyback co nverter is the turn on time point of the primary switch. In a har d switching mode the turning on of the MOSFET is not synchro nized with the drain-sourcevoltage value. This type of converters runs mainly in fixed

24、 frequencymode.電感電流斷續工作的反激式變換器一次側開關可工作于 硬開關或準諧振(或谷值開關或零電壓開關)模式。硬開關 和準諧振反激變換器之間的差異在于一次側開關的開啟時 間點。在硬開關里場效應晶體管的開啟波形拐點并不和漏源 極電壓值同步。這種變換器大體上運行于固定頻率模式。In a quasi resonantmodethe resonantcircuit determinedby t he output capacity of the MOSFET and the inductanceof the tr ansformerwill be utilized to switch

25、on at lowest possiblevalue o f the drain-sourcevoltage. This circuit starts to oscillate at the en d of the current flow through the secondaryside of the transform er, henceat the end of the flyback phase.The MOSFET will be turned on at the minimum of this oscillation. The quasi resonant approachuse

26、sthis oscillation to achieveminimum voltage switching during turn on for the MOSFET. This operation moderuns at a variable frequency.在準諧振模式里，由變壓器電感和場效應晶體管輸由電容 引起的諧振促使開關的開通時刻發生在漏源極電壓的最小 值上。這種電路在電流從變壓器二次側流盡以后（反激回掃 過程結束）開始振蕩。場效應晶體管將在振蕩幅值的最小值 開啟（谷值開通）。這種運行模式工作在可變的頻率上。Higher amplitude of the oscillation

27、results in lower drain sourc e voltage level at which the MOSFET turns on correspondingly wer switching lossesand higher efficiency of the system.更高幅值的振蕩導致場效應晶體管更低的漏源極開通電壓幅值來產生更低的開關損耗和更高的系統效率。To achievehigh oscillation peaks,the designof the transformer has to be set to high reflectedvoltage. This i

28、ncreaseof the refle cted voltage results in a higher drain-sourcevoltage blocking MOS FET and longer duty cycles.要達到比較高的振蕩電壓峰值，變壓器的反射電壓必須設置的比較高。增加的反射電壓導致使用更高漏源極擊穿電壓 的場效應晶體管和更大的開關占空比。Comparisonof three different flyback solutions has beenmade. All of them have beenoperation at 300 kHz, bus voltage of

29、 40 0 V, output power of 120 W, output voltage of 16 V. Thesede sign included different modesof operation and different values of reflectedvoltage, resultng in different MOSFET s voltage ratings:比較現有的三種反激變換器。它們都工作在3 0。千赫茲，直流母線電壓4 0 0伏特，輸由功率1 2 0瓦特，輸由電壓1 6伏特。這些設計包含不同的運行模式和反射電壓等級，因此使用不同電壓等級的場效應晶體管：Har

30、d switching flyback with CoolMOS 600V, reflectedvoltage of 100V硬開關反激變換器使用6 0 0伏特 CoolMOS 1 0 0伏 特反射電壓Quasi resonantflyback with CoolMOS600V, reflectedvoltage of 100V準諧振反激變換器使用6 0 0伏特 CoolMOS 1 0 0伏 特反射電壓Quasi resonantflyback with CoolMOS800V, reflectedvoltage of 390V準諧振反激變換器使用80 0伏特CoolMOS 390伏特反射電壓

31、The clampingsnubbercircuit was set to the rated breakdownvoltage of the MOSFET (600 V and 800 V respectively).鉗位緩沖電路被設定在場效應晶體管的額定擊穿電壓上（分別為6 0 0伏特和8 0 0伏特）Flyback in hard switching modewith 600V MOSFET 使用6 0 0伏特場效應晶體管的硬開關反激變換器The hard switching approach(as shown in Fig. 26) doesn 'ct onsiderthe m

32、inimum drain-sourcevoltage. The MOSFET will be t urned on hard, in this caseat a voltage level of 500 V (at time point 3.3 以 s)The dischargeof circuits parasitic capacitancesea ds to a high current spike during turning on.硬開關(圖2 6所示)幾乎不考慮漏源極電壓的最小值。 場效應晶體管開通應力大，在這個例子里，開通電壓在5 0 。伏特(在3 .3微秒的時間點)。由寄生電容引

33、起的泄放電 流在開通時產生很高的電流尖刺。Fig. 26 Drain-sourcevoltage and drain current of hard switching600V flyback圖2 66 0 0伏特硬開關反激變換器的漏源極電壓和漏極電流Flyback in quasi resonantmodewith 600 V MOSFET 使用6 0 0伏特場效應晶體管的準諧振反激變換器The drain-sourcevoltage (Fig. 27) starts oscillating at the end of the flyback phaseand reachingthe min

34、imum of 300 V when the MOSFET turns on.漏源極電壓(圖2 7)在反射過程結束后并減小到3 0 0 伏特時場效應晶體管導通。The duty cycle is lower comparedto an 800 V solution due to a lower reflectedvoltage of 100V. Shorter duty cycle for the sameoutput power results in higher peak currents on the primar y side.因為1 0。伏特的反射電壓，比較8 0 0伏特解決方案 它

35、有更小的占空比。小占空比實現同樣的功率輸生必須使用 更高的一次側峰值電流。Fig. 27 Drain-sourcevoltage and drain current of quasi resonant600V flyback圖2 76 0 0伏特準諧振反激變換器的漏源極電壓和漏極電流Flyback in quasi resonantmodewith 800 V MOSFET 使用8 0 0伏特場效應晶體管的準諧振反激變換器The drain-sourcevoltage (Fig. 28) starts oscillating at the end of the flyback phaseand

36、 reachingthe minimum of 100V when t he MOSFET turns on. The turning on current spike is low.漏源極電壓(圖2 8)在反射過程結束后并減小到1 0 0 伏特時場效應晶體管導通。開通電流尖刺比較低。The duty cycle is higher comparedo a 600V solution due t o a higher reflectedvoltage of 390V. Longer duty cycle for the s ame output power results in lower p

37、eak currents on the primary s ide.因為有3 9。伏特的反射電壓，所以有比6 0 0伏特解 決方案更大的占空比。更大的占空比實現同樣的輸生功率可 以使用更低的一次側峰值電流。O.E4OO l,E-06Fig. 28 Drain-sourcevoltage and drain current of quasi resonant800V flyback圖2 88 0 0伏特準諧振反激變換器的漏源極電壓和漏極電流Comparisoof spectra干擾頻譜比較The spectraof the drain-sourcevoltagesfor correspondinglyba ck design(Fig. 26Fig. 27 and Fig. 28) are shown in Fig. 29.相應設計的反激變換器(圖2 6、圖2 7和圖2 8)的漏源極電